The present invention is directed in general to wireline telecommunication systems, and more particularly to an enhancement of a frequency domain reflectometry (FDR)-based, energy reflection anomaly-locating mechanism of the type disclosed in the ""681 application, including the use of a precursor signal conditioning circuit for improving the performance of the FDR signal processing subsystem.
As described in the above-referenced ""681 application, telecommunication service providers are continually seeking ways to optimize the bandwidth and digital signal transport distance of their very substantial existing copper plant, which was originally installed for the purpose of carrying nothing more than conventional analog (plain old telephone service or POTS) signals. In addition to the inherent bandwidth limitations of the (twisted pair) copper wire medium, service providers must deal with the fact that in-place metallic cable plants, such as that shown at 10 in the reduced complexity network diagram of FIG. 1, linking a central office 12 with a subscriber site 14, typically contain one or more anomalies, such as but not limited to load coils (used to enhance the wireline""s three to four kilohertz voice response), and bridged taps 16, to which unterminated (and therefore reflective) lateral twisted pairs 18 of varying lengths may be connected.
Because these discontinuities cause a portion of the energy propagating along the wireline link to be reflected back in the direction of the source, at the high frequencies used for digital data communications (e.g., on the order of one MHz), such reflections can cause a significant reduction in signal amplitude, when (counter-phase) combined with the original signal, disrupting digital data service. To locate these reflection points, it has been conventional practice to employ interactive, time domain reflectometry (TDR), which relies upon the ability of a skilled technician to make a visual interpretation of a displayed TDR waveform, and thereby hopefully identify the bridged taps, and the lengths of any laterals that may extend therefrom. Because this process is subjective, it is imprecise and very difficult to automate.
In accordance with the invention disclosed in the ""681 application, shortcomings of a conventional TDR-based scheme for locating energy reflecting anomalies are obviated by stimulating the line with a linearly stepped frequency sinusoidal waveform, and analyzing the composite waveform response by means of frequency domain reflectometry, whose frequency bins represent distances that are integral multiples of delay, so that there is a one-for-one correspondence between the bins of a Discrete Fourier Transform (DFT) and distances to the reflection points along the wireline.
The frequency domain reflectometry system of the ""681 application is diagrammatically illustrated in FIG. 2 as comprising a processor-controlled test head 20 (such as may be installed in a central office, or included as part of test signal generation and processing circuitry of a portable craftsperson""s test set), coupled to an access location 21 of a line under test (LUT) 22 by means of a line-driver amplifier 24 and an input receiver amplifier 26. Line-driver amplifier 24 is coupled to the LUT 22 through source resistors 27, 28, each having an impedance equal to one-half the impedance (Zo) of the metallic line pair.
Coupled to the test head 20 is a control processor 30, that is programmed with an FDR test routine shown in the functional block diagram of FIG. 3. As shown therein, an initial tone generation function 31 generates a series of digitally created test signals, in particular a sequence of discrete frequency sinusoidal tones, to produce what is in effect a frequency-swept sinusoidal waveform. The swept frequency waveform may be varied in a linear, stepwise manner, for example beginning at minimum frequency such as 0 Hz and stepped in incremental frequency steps up to a maximum frequency. (Conversely, the frequency variation may begin at an upper frequency and proceed to a minimum frequency, without a loss in generality.) These tones are applied (via the line-driver amplifier 24 of FIG. 2) to the line under test 22.
As the frequency of the sinusoidal waveform is swept, the wireline""s response signal level at the test access point 21 is monitored (via the input amplifier 26), digitized by way of an analog-to-digital converter (ADC) 32, and stored in a signal measurement buffer (not shown). The amplitude of the measured signal response will exhibit a variation with frequency that is a composite of the fluctuations in impedance due to any reflection points along the LUT. In order to optimize the accuracy of the analysis, the response data may be selectively modified by a bandpass filter BPF 33, the center frequency of which is varied, or xe2x80x98slidesxe2x80x99, along the variation of frequency of the swept sinusoid being applied to the LUT. This filtering operation serves to remove any DC level and discontinuities that might cause spurious results, between start and end sample values of the data. The filtered data is then stored with each frequency step iteration, to produce a sampled amplitude array 34. A loss compensation function (LCF) may also be applied to the data set, to compensate the frequency response characteristic of the LUT for loss over distance and frequency.
The line under test can be characterized in terms of its resistance (R), inductance (L), capacitance (C), and conductance (G) parameters per unit length, which are available from tabulated industry-available sources for the type of wire. From these parameters, a frequency-dependent propagation constant xcfx84 can be derived as:
xcfx84=xcex1+jxcex2=((R+jwL)(G+jwC))1/2, where w=2Πf.
The real part of the propagation constant, xcex1(f), is the attenuation along the line per unit length. Since the envelope of a signal propagating along the line as a function of distance is attenuated by exe2x88x92xcex1(f)t, xcex1(f) can be determined.
The effect on the frequency response waveform is that amplitude decay is less pronounced for reflected signals propagating on shorter loops, since the shorter distance offsets the effects of the loss at high frequencies, due to the effects of xcex1(f). Since the actual length of the line under test is unknown, a compromise between the two extremes may be employed, to provide compensation for the overall frequency response waveform for all distances of interest.
In order to determine the coefficient of the exponential attenuation function in terms of frequency, it is necessary to reduce the number of degrees of freedom of the total loss function. Since the maximum frequency of the swept sinusoidal waveform is known, a priori, a loss compensation function based upon the mid frequency point of the sweep fmid=fmax/2 may be employed. As will be described in detail below with reference to the amplitude vs. frequency response diagram of FIG. 4, from this mid frequency, fmid, a corresponding resolution distance dmid is defined as:
dmid=Vp/4fmid
An xe2x80x98average lossxe2x80x99 value xcex7 can be derived as:
xcex7=exe2x88x92xcex1(fmid)dmid
The loss compensation function LCF can therefore be defined as:
LCF=exp((xe2x88x9221n(xcex7)/fmax)f).
This loss-compensated data is processed in accordance with a frequency analysis operator, such as Discrete Fourier Transform (DFT) 35, which decomposes the composite line signal response into frequency bins associated with the individual reflectors"" frequency fluctuations.
Next, the Fourier transform-processed data is coupled to a tap decision operator 36, the output of which is coupled to a remote terminal unit 37. The tap decision operator employs a threshold established for the contents of the frequency bin data produced by the DFT, in order to distinguish between significant (useful) and spurious energy. The threshold employed may be is defined as:
Threshold(bin no.)=[(StartValuexe2x88x92EndValue)*exp(xe2x88x92bin no.*slope)]+EndValue.
The parameters StartValue, EndValue and slope are dependent upon the test head circuitry""s gain and swept bandwidth, and are empirically determined. A frequency bin is considered to contain significant energy, if its contents exceed the threshold for that bin number.
Any frequency bin whose contents exceed its threshold are subjected to waveform analysis of the type used in frequency domain reflectometry. For an arbitrary waveform v(t) that is the sum of two waveforms of some frequency fo, a minimum will occur in v(t) at some delay to of one waveform relative to the other. For a wireline cable plant, this occurs when a waveform vo propagating downstream along the wireline is combined with a waveform v1 reflected from an anomaly, such as a bridged tap and returning upstream along that wireline.
In general, the combination of these two waveforms can be expressed as:
v(t)=vo(t)+v1([txe2x88x92to])
v(t)=Vo sin (2Πfot)+V1 sin (2Πfo[txe2x88x92to]).
Since the downstream and upstream propagating waveform components have the same frequency, v(t) will have a local minimum due to destructive interference at some time delay to when the arguments of vo and v1 differ by Π radians. Namely,
xe2x80x83(2Πfot)xe2x88x92(2Πfo[txe2x88x92to])=Π.
Dividing this expression by 2Πfot and solving for to, yields:
to=1/2fo=To/2,
where the period To of the waveform is 1/2fo.
As shown in the decaying waveform amplitude vs. frequency diagram of FIG. 4, a null occurs at fo; nulls in v(t) also occur for frequencies fk, where fk greater than fo, and the arguments of vo and v1 differ by odd multiples of Π. If k is a positive integer, the nulls will occur when:
(2Πfkt)xe2x88x92(2Πfk[txe2x88x92to])=2Πfkto=(2k+1)Π.
Letting the period Tk=1/fk, then
2Πfkto=2Πto/Tk=2Π(To/2)/Tk=(2k+1)Π.
To/Tk=(2k+1)
Substituting To=1/fo, Tk=1/fk:
fk/fo=(2k+1),
or
fk=fo(2k+1), for k=0, 1, 2, . . . 
The periodicity of the nulls can be seen by examining the difference in frequency between two adjacent nulls fm and fm+1.
From the foregoing, fm+1xe2x88x92fm=fo(2[m+1]+1)xe2x88x92fo(2m+1)=2fo, for m=0, 1, 2, . . .
This means that a linear sweep of a wireline having a single reflection point (e.g., bridged tap) will produce nulls in the frequency response at frequencies fo, 3fo, 5fo, 7o, etc., as shown in the amplitude vs. frequency response diagram of FIG. 4.
Denoting Fo as the repetition rate of the nulls for t=to in the frequency domain, then:
xe2x80x83Fo=1/(period of the null)=1/(fm+1xe2x88x92fm)=xc2xdfoΠ.
In general, the null repetition rate in the frequency domain Fn is given by: Fn=1/2fn, where fn is the lowest frequency at which a null occurs when the delay t=tn.
From the above relationships, Fo corresponds to to and, in general Fn corresponds to tn, and is the same as the round-trip delay of the signal from the wireline access location to the point of reflection and back. In order to determine the length of time required for the waveform to propagate to an impedance-mismatch reflection point, it may be observed that to is representative of the total time required for the downstream propagating waveform to be reflected back to the access location 21 at which the measurement is taken. This one-way delay ti=to. To determine the distance to this reflection point from the access location, the propagation velocity vp of the waveform along the wireline must be known.
In general, using xcex5r as the dielectric constant of the wireline insulation, c as the velocity of light in free space, and xcexcr as relative permeability, then the propagation velocity along the wireline may be expressed as: vp=c(xcex5rxcexcr)xe2x88x921/2.
Knowing the type of cable from industry available specifications allows the propagation velocity (typically on the order of ⅔ the velocity of light) to be readily determined.
The distance D from the access location to the location of the impedance mismatch reflection (e.g., bridge tap) may be given by the expression:
xe2x80x83D=tivp=vpto/2=Tovp/4.
Thus, D is proportional to To/4, which is one-quarter wavelength of the sinusoid waveform having a frequency fo. Substituting To=1/fo, yields D=vp/4fo.
Namely, the distance D is inversely proportional to frequency. This means that the minimum resolvable distance Dmin=vp/2fmax.
As pointed out above, the response waveform v(t) seen at the signal measurement point will contain components produced by a plurality of reflection points as:
v(t)=vo(t)+v1(txe2x88x92to)+v2(txe2x88x92t1)+v3(txe2x88x92t2)+ . . . vn(txe2x88x92tnxe2x88x921).
Since these components are associated in general with impedance discontinuities caused by physical characteristics in the wireline separated by varying distances from the source, the delays to, t1, . . . tnxe2x88x921, associated with these reflections will be mutually different, so that the values To/2, T1/2, . . . Tnxe2x88x921/2, and thus the frequencies fo, f1, . . . fnxe2x88x921, will be mutually different.
As fn is unique for each delay, then by identifying the various frequencies fn, the two-way delay times tn of a reflection from a wireline discontinuity may be readily determined. As pointed out above, once the time delay is known, the distance D to the impedance mismatch discontinuity (e.g., bridged tap) may be readily determined.
To determine the individual values of two-way delay time tn, a frequency response waveform a(f) produced by stimulating the wireline with a linearly swept sinusoidal waveform is formed of samples at discrete frequency steps of (fmax/N). For a radix-two buffer size of N points, the output of the DFT operation will yield values that are proportional to the magnitudes of the various null repetition rates Fk. If the maximum frequency fmax of the swept sinusoid waveform is 2fo, then the minimum resolution of the DFT is:
Minimum resolution=1/fmax=1/2fo=Fo=to (seconds).
Denoting the contents of frequency bin m as A(m) of the DFT of a(k), then the contents A(1) of the first frequency bin are the DC component of the swept response, while the bin m contains the magnitude of the null repetition rate (mxe2x88x921)Fo, for m=2, 3, 4, . . . N/2. Namely, the various energy bins of the response A represent the energy in a(f) associated with the different round trip time delays to, 2to, 3to, etc., and A(m) contains the magnitude of the waveforms delayed by (mxe2x88x921)to for m=2, 3, 4, . . . N/2. The contents of the bins are used to calculate distances from the wireline access location to the energy-reflecting anomalies. In particular, the distance DRP to a reflection point RP is determined by multiplying the one-way delay tRP by the velocity of propagation vp of the waveform. Namely, Dmxe2x88x921=(mxe2x88x921)tovp/2=[(mxe2x88x921)to]vp/2 for m=2, 3, 4, . . . N/2, so that there is a one-to-one correspondence between the bins of DFT and the distances to the reflection points along the wireline.
Although the FDR scheme described above works well for short to medium distance lines (e.g., up to wireline distances on the order of 15 Kft), it has been found that the signal-to-noise ratio (SNR) of the processed response characteristic decreases dramatically for longer distances (e.g., on the order of 18 Kft and beyond).
In accordance with the invention, this longer distance-associated SNR reduction problem is successfully addressed by modifying the FDR processing mechanism disclosed in the ""681 application, to incorporate a prescribed precursor response filtering operator prior to the Fourier processing operation. As will be described, this precursor response filter may take the form of a precise curve-fitting operator or a piecewise high pass filter bank. In addition, the Fourier processing operator is implemented as a Fast Fourier Transform (FFT) operator rather than a Discrete Fourier Transform (DFT). The FFT operator may in actuality comprise an inverse FFT operator (FFTxe2x88x921), since the input to this block is frequency data and its output is related to time (distance) data, and performing either of an FFT or an inverse FFT on the input data produces identical outputs.
The xe2x80x98best-fitxe2x80x99 curve-based precursor response filter of the first embodiment produces a response characteristic which very closely xe2x80x98fitsxe2x80x99 the decaying profile of the amplitude array along which ride the very small perturbations. This best fit profile is differentially combined with the original amplitude array to realize a set of more clearly delineated amplitudes along the swept frequency band, which are readily identified in the FFT to which the filtered data is applied.
The high pass filter bank of the second embodiment is formed of a pair of parallel high pass filters having Z-domain transfer functions associated with respectively different distance wireline segments. One of the filters detects discontinuities at relatively short to medium distances (e.g., on the order of distances up to 10 Kft from the wireline access location) associated with relatively low frequency components, while the other high pass filter detects discontinuities at relatively greater distances (e.g., on the order of distances from 10 Kft to 18 Kft and beyond from the wireline access location) associated with relatively high frequency components.
The outputs of the high pass filters are respectively coupled to associated FFT operators whose outputs are piece-wise combined to realize a composite characteristic that is effective to emphasize response components over the total distance of the two distance segments. A level-shifter may be used to provide for a relatively smooth coupling of the two FFT response characteristics.
Although each of the above embodiments enhances the ability of an FDR system of the type to extract very small amplitude signals particularly those of the higher frequency tones, and thereby locate energy reflection anomalies such as bridged taps along a relatively extended distance wireline, the ability to process such reduced amplitude response signals means that the signal processing system must have a very large dynamic range. This implies that the resolution of the analog-to-digital converter used to digitize the data values of the amplitude array must be relatively wide.
This dynamic range issue is handled by installing a signal conditioning circuit between the test head and the analog-to-digital converter. The signal conditioning circuit is comprised of a cascaded arrangement of a comb filter bank contains N band-pass filters, an envelope detector, and a compander. The envelope detector converts the output frequencies from the comb filter into DC levels corresponding to the amplitudes of the peaks of the tones. The compander increases the gain of higher frequencies, so that, given an exponentially decreasing input function, it produces a relatively linear output, with a more uniform representation in the input range of the analog-to-digital converter, as desired.